Detector system with adjustable phase stage

ABSTRACT

A detector system has a local oscillator port, a radio frequency port, and an oscillator providing a local oscillator signal having amplitude-modulated (“AM”) noise coupled. A first detector detects the local oscillator signal and produces a first detected signal having a first detected AM noise signal component and a demodulated signal component at a first phase. A second detector detects a second high-frequency signal to produce a second detected signal having a second detected AM noise signal component at a second phase. An algebraic combining network combines the first detected signal and the second detected signal to produce a demodulated signal component at an output. A phase delay disposed between the local oscillator and the output of the algebraic combining network is calibrated to offset the first phase from the second phase to improve signal-to-noise ratio at the output.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application is a continuation patent application of U.S.patent application Ser. No. 12/590,145 entitled NOISE-CANCELING DOWNCONVERTING DETECTOR by Moulton et al., filed Nov. 2, 2009, the contentsof which are hereby incorporated by reference in their entirety for allpurposes, and claims the benefits of U.S. patent application Ser. No.12/590,145 under 35 U.S.C. 120.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

None.

NAMES OF PARTIES TO A JOINT RESEARCH AGREEMENT

None.

REFERENCE TO A SEQUENCE LISTING

None.

FIELD OF THE INVENTION

This invention relates generally to electromagnetic field disturbancesensing techniques, such as radar systems.

BACKGROUND OF THE INVENTION

Continuous Wave (CW) Coherent Radar uses frequency conversion to comparethe phase of a transmitted signal with the reflection of that signalfrom a moving target. The phase of the wave reflected from the targetchanges as a function of the changing distance to that target. If thetarget velocity remains constant, the phase of the reflected signalchanges at a constant rate. A constant rate of phase change correspondsto a constant frequency. Thus the returning reflected signal appears ata frequency offset from the transmitted signal that is proportional tothe relative velocity between the transmitter and the target.

Comparing the transmitted signal and the received signal with afrequency downconverter delivers the difference frequency between thetransmitted and received signals at the converter output. Practicalimplementation of a radar seeks to optimize the cost and size of theequipment required to compare the phase (or frequency) of thetransmitted and received signals, while obtaining the greatest detectionrange to target possible for that cost and size.

Many conventional portable radar guns use a Gunn diode driving a cavityoscillator with an integral diode peak detector which functions as afrequency downconverter or mixer, using either one or multiple detectordiodes. The cavity oscillator/mixer is coupled to a horn antenna used totransmit the incident signal and to receive the reflected signal. Thecavity drives the diode detector with a local oscillator (“LO”) signalfrom the transmitter and couples the received RF signal to the samediode. The diode detector mixes the RF and LO signals, creating an IFsignal at their difference frequency. The diode detector typicallymatches to a relatively high impedance, often hundreds or even thousandsof ohms, and conversion loss can approach 0 dB. Matching to LO and RFsignals is accomplished by moving the diode location within the cavityto optimize the coupling for optimal system performance.

The detector diode also rectifies the LO power in the cavity, and anyvariations in the amplitude due to either coherent amplitude modulation(“AM”) or to AM noise will show up at the IF output. Because of thisproblem, designers typically use Gunn diode oscillators adjusted to thepoint of minimal conversion of diode bias supply voltage input toamplitude variation. This minimizes the AM noise on the LO and thus alsominimizes the detected LO AM noise on the IF output allowing forsufficiently sensitive RF detection.

The cavity based radar devices typically require a horn antenna up toseveral inches long and a cavity oscillator at least one or more cubicinches in size for operation at the 10 GHz or 24 GHz ISM bands (e.g.,the X, and K bands). Both of these factors cause the system to havesignificant weight and size, which is undesirable for a small hand-heldapplication. Furthermore, the optimum Gunn diode bias point oftenrequires substantial current draw, limiting the useful operating timefor portable, battery-powered applications. Alternatively, the radarsize must increase to accommodate larger batteries.

Another design approach to small sized radar devices uses planar or“patch” antenna arrays. These devices either use cavity stabilized Gunnoscillator/detectors or use traditional switching mixers where the LOsignal switches the RF signal phase to the IF output dependent upon LOphase. The switching type of mixer typically shows 6 dB or moreconversion loss, and must be a balanced configuration to cancel any AMnoise of the local oscillator. Diodes used in conventional mixer-basedsystems act like switches that provide either an open circuit or aclosed switch in a signal path. The LO signal drives the mixer diode(s)to turn the diode “on”, or low impedance, for about a half cycle and“off”, or high impedance, for the other half cycle.

The balanced or double or triple balanced switching diode mixer suffersfrom imperfect AM noise cancelation due to variations in manufacturingand remains sensitive to the AM noise of most oscillators. Thedown-converted local oscillator AM noise obscures the incoming RFsignal, even while the local oscillator phase noise cancels due to theshort time required for the round trip on the radar path or the pathinside the mixer itself. Conventional (incoherent) receivers do nottypically see the AM noise of the LO as the phase noise typicallydominates the AM noise by tens of dB. Only in coherent reception (suchas used for CW radar) does the phase noise of the LO cancel and allowthe AM noise to dominate.

Additionally, the IF output of a switching diode mixer typicallyrequires termination with a low noise IF amplifier with low inputimpedance, usually equal to 50 ohms. The noise voltage of that amplifierwith 6 dB mixer loss is equivalent to twice that noise voltage measuredat the antenna input. Diodes typically add another 0.5 to 1 dB to theinput noise of the mixer above the conversion loss, further degradingthe receive signal to noise ratio as seen at the antenna RF port. Thistype of radar does not typically deliver good long range performancecompared with the Gunn and horn antenna alternatives without theaddition of other components such as additional antennas or an RFpreamplifier.

Other devices constructed using planar patch antenna arrays have used aGunn-based cavity oscillator for the transmitter source and a detectordiode for the receive mixer. These can provide reasonable AM noise fromthe Gunn source, but are limited in miniaturization by the size of theoscillator resonant cavity.

Components for radar systems and other applications overcoming thedeficiencies of the prior art are desirable.

SUMMARY OF THE INVENTION

A detector system has a local oscillator port, a radio frequency port,and an oscillator providing a local oscillator signal havingamplitude-modulated (“AM”) noise coupled to the local oscillator port. Afirst detector configured to detect at least the local oscillator signalproduces a first detected signal having at least a first detected AMnoise signal component and a demodulated signal component at a firstphase. A second detector configured to detect a second high-frequencysignal having the AM noise produces a second detected signal having atleast a second detected AM noise signal component at a second phase. Analgebraic combining network combines the first detected signal and thesecond detected signal to produce an output signal including thedemodulated signal component at an output. A phase delay disposedbetween one of the local oscillator port and the radio frequency portand the output of the algebraic combining network is calibrated tooffset the first phase from the second phase to improve signal-to-noiseratio at the output.

In a particular embodiment, the local oscillator signal is at a selectedlocal oscillator signal level and the phase delay is calibrated at theselected local oscillator signal level. A further embodiment has ahigh-frequency combining network and the phase delay is disposed in aradio frequency path of the high-frequency combining network. In analternative embodiment, the phase delay is disposed in a localoscillator path of the high-frequency combining network.

In another embodiment, the phase delay is disposed between the firstdetector and the output of the algebraic combining network. In analternative embodiment, the phase delay is disposed between the seconddetector and the output of the algebraic combining network. A furtherembodiment includes a second phase delay calibrated to improvesignal-to-noise ratio at the output.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a noise canceling down-converting detectoraccording to an embodiment.

FIG. 2A is a block diagram of a noise canceling down-converting detectoraccording to another embodiment.

FIG. 2B is a block diagram of the noise canceling down-convertingdetector of FIG. 2A showing additional details of the high-frequencycombining network and the detected signal algebraic combining network.

FIG. 3A is a diagram of a portion of noise-canceling system according toan embodiment.

FIG. 3B is a diagram of a portion of a noise-canceling system accordingto another embodiment.

FIG. 4 is a diagram of portion of a noise canceling down-convertingdetector system according to an embodiment.

FIG. 5A is a circuit diagram of a single-diode detector used in anembodiment.

FIG. 5B is a circuit diagram of a dual-diode detector used in anembodiment.

FIG. 5C is a circuit diagram of a diode multiplier circuit used in anembodiment.

FIG. 5D is a circuit diagram of a single-diode detector in a shuntconfiguration used in an embodiment.

FIG. 6A is a plan view of a noise-canceling down-converting detector RFcoupler for use in a field disturbance sensing system according to anembodiment.

FIG. 6B is a plan view of a single-diode detector circuit according toan embodiment.

FIG. 6C is a plan view of a dual-diode detector circuit 660 according toan embodiment.

FIG. 7A is a plan view of a circuit board of a CW radar system accordingto an embodiment.

FIG. 7B is a diagram of a field disturbance measuring system accordingto an embodiment.

FIG. 8A is flow chart of a method of down-converting according to anembodiment.

FIG. 8B is a flow chart of a method of sensing an electro-magnetic fielddisturbance according to an embodiment.

FIG. 9A is a flow chart of a method of calibrating a field disturbancesensing system for noise according to an embodiment.

FIG. 9B is a flow chart of a method of calibrating AM signal gain of afield disturbance sensing system according to an embodiment.

DETAILED DESCRIPTION OF THE DRAWINGS

Field disturbance sensing systems (e.g., radar systems, motion sensors,automatic door openers, automotive sensors, and low-IF Tx/Rx systems)according to embodiments achieve small size and improved range,sensitivity and signal-to-noise ratio. Systems according to someembodiments are mobile, battery-powered systems. Other embodiments arefixed installations with AC (mains) power or external power supplies. Anoise-canceling down-converting detector according to an embodimentcancels the AM noise from a first signal (signal 1, which will bereferred to as an “LO signal”, although it performs quite differentlyfrom an LO signal in a switching mixer system) or transmitter signal.

Some embodiments of detectors provide significant conversion gain whenproperly matched to the detector circuit compared to a conventionalswitching mixer-based system. In some embodiments, greater than 10 dB ofconversion voltage gain has been measured. Improved conversion gainallows use of a simple, low cost IF amplifier as an alternative to themore expensive low noise IF or RF amplifiers used in conventionalswitched diode mixer systems because of the superior signal to noiseratio embodiments achieve. In some embodiments, low noise IF amplifiersare used to boost the converted signal with minimal additional noise. Asused herein, “noise canceling” means AM noise detected at one detectoris subtracted from essentially the same AM noise detected at anotherdetector.

A noise canceling down-converting detector according to an embodimentdetects an incoming (reflected or other RF) signal while canceling AMnoise on the local (transmitted or LO) signal. The noise cancelingdown-converting detector does not operate as a traditional switchingmixer in that it does not switch or gate the RF signal to an IF portwith the LO signal controlling the switching action. Instead, the noisecanceling down-converting detector operates as two or more separateenvelope detectors.

The detectors detect the instantaneous value (voltage) of twohigh-frequency signals. One of these signals includes the average(steady state) LO power (generating an essentially DC detected output),the instantaneous RF power, which adds to or subtracts from the LOsignal to produce a beat (IF) signal, and the instantaneous LO AM noisepower, which can also add to or subtract from the average LO power, thusintroducing inaccuracy into the peak detected voltage (potentiallyobscuring or adding uncertainty to the detected RF signal). Thesignal-of-interest (e.g., the reflected signal or a received signal froma transmitter) is demodulated into a component of the detected signal(“demodulated component”).

The other signal includes at least the average LO power and theinstantaneous LO AM noise power (and optionally the RF signal or othersignals). The detected steady-state LO signals are DC, and are rejectedor otherwise canceled (e.g., do not affect an RC detector load). Thedetected AM noise components are subtracted from each other, thuscanceling the effect of LO AM noise on the resultant detected (IF)signal. The RF signal, which in some embodiments is the reflected signalfrom a moving object, adds to or subtracts from the instantaneous LOsignal at the detector and increases or decreases the peak voltage, andhence the detector output. A convenient visualization is that thereflected signal is “sliding past” the LO signal, and the detectorproduces an IF signal at a beat frequency in base band.

In conventional switching mixers, the IF signal is the sum or differencebetween two frequencies (i.e., the RF and LO signals). In other words, aconventional switching mixer can operate as an upconverter or adownconverter. Conventional switching mixers typically operate in arelatively low impedance system (e.g., a system with a characteristicimpedance of fifty ohms) at all mixer ports; LO, RF and IF. Noisecanceling down-converting detectors are not limited to systems with suchlow impedance at the IF port.

FIG. 1 is a block diagram of a noise canceling down-converting detector(“NCDD”) system 100 according to an embodiment. A high-frequency signal(“LO signal”) 102 is provided to a first detector 104 and to analgebraic combining network 106 that combines the high-frequency signal102 with the RF signal 108 to produce an RF+LO signal 110, which isprovided to a second detector 112. In some applications, such as CWradar, the high-frequency signal is a transmitted signal and RF signalis a reflected signal, namely the reflected LO from a target (see, e.g.FIG. 7B). In other applications, the RF signal is a high-frequencysignal that is generated by a transmitter device, or a re-generated LOsignal. While many applications of embodiments use a coherent RF signal,other applications do not.

In a continuous-wave (“CW”) radar system, a high-frequency (LO) signalis sent to an antenna and transmitted to a target, which reflects aportion of the energy (the “reflected” or “RF” signal) that is receivedby the same antenna or by a second antenna. Relative motion between thetarget and antenna(s) is detected as a frequency shift between thereflected signal and the transmitted (LO) signal. The detected frequencyshift is processed to determine the speed of the relative motion. The LOsignal is typically much higher power than the reflected signal, and AMnoise on the LO signal can obscure the RF component.

The outputs 114, 116 from the detectors 104, 112 are provided to anon-inverting input 117 and an inverting input 115 of a second algebraiccombining network 118 to produce a detected output signal 120, whichwill be referred to as an intermediate frequency (“IF”) signal forpurposes of convenient discussion. The second algebraic combiningnetwork 118 can be implemented in any of several types of circuits, suchas an operational amplifier, differential amplifier, or a digitalcircuit technique incorporating analog-to-digital converters andsubsequent digital signal processing in hardware or software. The secondalgebraic combining network combines the first detected signal and thesecond detected signal to cancel the detected AM noise. The firstdetected signal 116 includes a detected AM noise component (LO_(d)) anda second detected signal component (RF_(d)). The second detected signal114 includes a detected AM noise component (LO_(d)) nominally equal tothe detected AM noise component in the first detected signal 116, thus,the detected AM noise cancels. The output signal (IF) 120 is the seconddetected signal component (RF_(d)).

The IF signal of FIG. 1 is basically the demodulated RF signal, which isessentially a beat note as the phase of the RF signal advances orretreats in relation to the CW LO signal. In the case of a CW radarsystem, the phase of the RF signal advances or retreats relative to thephase of the LO signal according to the speed of an object reflectingthe RF signal back to the system as the object moves toward or away fromthe LO source and the path length to and from the object decreases orincreases. The period of the beat note indicates the speed of theobject. In an ideal system, the detected LO signal amplitude (DCcomponent of the detected LO signal) is the same at both detectors 112,104 and cancels out at the inputs 115, 117 of the algebraic combiningnetwork 118. Alternatively, the DC components are not equal, but areblocked (e.g., capacitively), rejected, or otherwise do not affect thebeat frequency between the RF and LO signals.

The IF signal 120 equals the detected LO signal at input 117, minus thedetected LO signal at input 115, plus the detected RF signal at input117. The system is setup (e.g., matched design or calibrated) such thatthe output signals at 114 and 116 are equal in amplitude and phase for agiven incident power at the LO port 102, thus canceling the LO AM noiseon the signal output. In some systems, the amplitude of the LO signal ismuch higher than the RF signal and the LO AM noise can dominate thedifferences in the peak signal voltage arising from the RF signal. Thesystem 100 avoids the problem of LO AM noise dominating the RF signal bysubtracting the detected LO signal and associated LO AM noise from adetected copy of that same signal.

In a particular embodiment, the first and second detectors are singlediode detectors. In alternative embodiments, the first and seconddetectors are multiple diode detectors or other types of detectors. In aparticular embodiment the first and second detectors are substantiallyidentical to each other so that the detected LO signals aresubstantially identical and provide good cancelation of the LO AM noise.Manufacturing tolerances can result in minor differences, and someembodiments include calibration techniques, as discussed below inreference to FIG. 3B. In alternative embodiments, the detectors are notidentical, but are balanced. In other words, each detector producessubstantially identical output signals from the same input signal(s).Many variations in components and circuits are used in alternativeembodiments, as designing substantially identical circuits and usingsubstantially identical (i.e., the same part number, and in some cases,matched parts) is only one of many ways to achieve the desired signalbalance.

FIG. 2A is a block diagram of a noise canceling down converting detector200 according to another embodiment. The RF signal 108 and LO signal 102are both provided to a algebraic combining network 202 that produces afirst high frequency output 204 of LO+RF, and a second high frequencyoutput 206 of LO−RF. A first detector 208 detects the LO+RF signal andthe detected signal 216 (LOd+RFd) is provided to the non-inverting input217 of another algebraic combining network 218. The detected signal 214(LOd−RFd) from a second detector 210 is provided to the inverting input215 of the algebraic combining network 218. The output of the algebraiccombining network 218 is the IF signal, which is equal to the detectedLO signal from 217, minus the detected LO signal from 215, plus the sumof the magnitudes of detected RF signals seen at 215 and 217. When thedetected LO and RF signals are optimally combined, the detected LOsignal (including the detected LO AM noise) cancels, and the detected RFsignal doubles. This provides an IF signal with improved signal-to-noiseratio. In practice, differences arising from manufacturing tolerancesand from electrical components (e.g., detector diodes) result in slightimbalances that lead to residual LO AM noise and less than RF powerdoubling; however, significant performance improvement (voltage gain) ofthe conversion process is obtained.

FIG. 2B is a block diagram of the noise canceling down convertingdetector 200 of FIG. 2A showing additional details of the high-frequencycombining network 202 and the detected signal algebraic combiningnetwork 218. In a particular embodiment, the high-frequency combiningnetwork 202 includes a ring coupler (see, e.g., FIG. 6A). In alternativeembodiments, the high-frequency combining network 202 uses othertechniques such as, hybrids, baluns or transformers, or other combiningtechniques known to one skilled in the art of high-frequency circuitdesign.

The RF signal path (“first RF path”) 220 to the first detector 208 has afirst associated phase delay φ1 and a first gain (or loss, which will beexpressed as negative gain (or −dB) for purposes of discussion) g1, andthe RF signal path (“second RF path”) 222 to the second detector 210 hasa second associated phase delay φ2 and second gain g2. The LO signalpath (“first LO path”) 224 to the first detector 208 has an associatedphase shift φ3 and gain g3, and the LO signal path (“second LO path”)226 to the second detector 210 has a phase shift φ4 and gain g4.

The RF and LO signals are combined in high-frequency combiners 228, 230,and coupled to the detectors 208, 210. The detectors 208, 210 providelow-frequency signals to outputs 236, 238, and to the detected signalalgebraic combining network 218. The detected signals are subject tophase delay φ5, φ6, and gain g5, g6, before being combined in thelow-frequency summer 240 that adds the detected signals to produce acombined IF output 242.

Differences in the phase shifts and gains in complementary paths (i.e.,the first and second RF paths 220, 222 and the first and second LO paths224, 226) arising from component variation and manufacturing tolerancesin the combining network 202 can result in different high-frequencyoutput signals 232, 234 being supplied to the detectors 208, 210.Furthermore, differences in the detector components can result indifferent detector outputs 236, 238, even if the combined high-frequencysignals are equal. In some embodiments, one or more of the gain valuesin the high-frequency combining network 202 is adjustable. In aparticular embodiment, a gain value of an LO signal path (e.g., g3, g4)is adjustable to balance the detected LO power from each detector 236,238, which allows nearly complete cancelation of the detected LO AMnoise.

FIG. 3A is a diagram of a noise-canceling system 300 according to anembodiment. The detected signals LO_(d)+RF_(d) are provided to algebraiccombining network 302, having a positive output 306 and a negative(inverted) output 308. The positive output 306 couples LO_(d)+RF_(d) tothe first non-inverting input 310 of the low-frequency algebraiccombining networks 318 and the negative output 308 couples−(LO_(d)+RF_(d)) to a first inverting input 312 of the low-frequencyalgebraic combining networks 318. Similarly, LO_(d)−RF_(d) is coupled tothe second non-inverting input 314 and −(LO_(d)−RF_(d)) is coupled tothe second inverting input 316. The algebraic combining network 304operates similarly on LO_(d)−RF_(d) from the second detector.

The added common mode noise at the either differential outputs 306, 308or differential outputs 307, 309 will cancel at the output 320. Noisemay enter the differential output sum from power supply rails or fromother inputs. The system 300 cancels the common mode noise; however, itdoes not cancel differential noise showing up between outputs 306 and308, or between outputs 307 and 309.

FIG. 3B is a diagram of a portion of a noise-canceling system 330according to another embodiment. The output of the detectors 332, 334have different LO_(d) levels, which can arise due to different pathlosses or detector performance, for example. The RF_(d) levels may alsonot be equal; however, this is less critical as long as the totaldetected RF signal is sufficient to provide a valid measurement in aradar or other system because the radar determination of speed onlydepends upon the reliable detection of the beat frequency and not uponthe absolute amplitude of that signal. Different detected LO signalswould result in imperfect LO AM noise cancelation because the noisesignals would not be equal, which could deliver a higher detected AMnoise than the RF signal peak obscuring the accuracy of the RF signalmeasurement. The output from detector 334 is higher than the output fromdetector 332 by the scaling (gain) factor K. An adjustable gain stage336 multiplies the differential outputs by 1/K to drive the two detectedLO signals to the same level seen at detector 332, which results incancellation of the detected LO AM noise by the operation of thealgebraic combining network 330:

$\begin{matrix}{{IF} = {\left( {{LO}_{d} + {RF}_{d}} \right) + \left( {{- \left( {{LO}_{d} - {\left( {RF}_{d} \right)\text{/}K}} \right)} - \left( {{LO}_{d} - {\left( {RF}_{d} \right)\text{/}K}} \right) - \left( {- \left( {{LO}_{d} + {RF}_{d}} \right)} \right)} \right.}} & \left( {{Eq}.\mspace{14mu} 1} \right) \\{= {{LO}_{d} - {LO}_{d} - {LO}_{d} + {LO}_{d} + {RF}_{d} + {\left( {RF}_{d} \right)\text{/}K} + {\left( {RF}_{d} \right)\text{/}K} + {RF}_{d}}} & \left( {{Eq}.\mspace{14mu} 2} \right) \\{= {{\left( {2 + {2\text{/}K}} \right){RF}_{d}} + {2{LO}_{d}} - {2{LO}_{d}}}} & \left( {{Eq}.\mspace{14mu} 3} \right) \\{= {\left( {2 + {2\text{/}K}} \right){RF}_{d}}} & \left( {{Eq}.\mspace{14mu} 4} \right)\end{matrix}$

The gain control 336 allows canceling the LO AM noise, while increasingthe detected RF signal by approximately a factor of four (when the twodetected LO signals are approximately equal, i.e., when K is close tounity).

In a particular embodiment, a downconverting system is calibrated toachieve cancelation of LO AM signal or noise by adding an amount of AMsignal to each LO signal path of the system. By changing the gain K tocreate identical levels of detected LO AM signals at both detectoroutputs, the detected AM signal or noise seen at the IF will drop inamplitude. The LO AM canceling is done at baseband (IF/audio), whichallows for very precise LO AM calibration/cancelation compared tocalibrating at high frequencies (i.e., before detection), where mismatcherrors degrade the calibration accuracy. In a particular embodiment, adownconverting system has a built-in calibration source, such as an AMsignal or noise source (calibration standard), look-up table (“LUT”), orvariable gain or attenuation stage. In a further embodiment, thedownconverting system performs an automatic LO AM noise calibrationaccording to firmware instructions (self-calibration). LO AM noisecalibration provides improved signal-to-noise performance, which canprovide superior range to a radar system.

FIG. 4 is a diagram of portion of a NCDD system 400 according to anembodiment. The NCDD system 400 is similar to the systems described inreference to FIGS. 3A and 3B, hence a brief description is provided. Thesystem 400 uses four detectors D1, D2, D3, D4 providing four detectedsignals (as labeled in FIG. 4A) to the differential algebraic combiningnetworks 402, 404. The resultant combined IF output for the case of adifference circuit and equal LO and RF detected signals is:

IF=8RF _(d)+4LO_(d)−4LO_(d)=8RF _(d)  (Eq. 5)

The detectors D1 through D4 may each see as much as half of the incidentpower to the NCDD, if they are configured as dual diode detectors. Theydetect peak voltage and add their outputs as voltage. If they are pairedas a positive and negative peak detector, their low frequency detectedoutput voltage may nearly double for the same power input. Thus, thefour-detector system 400 cancels the detected LO AM noise whileincreasing the detected RF signal several times over what would bedetected by a single-diode system. Uncorrelated diode-generated (i.e.,detector generated) noise adds as power. In single diode detectorsystems, this uncorrelated noise is added to the detected signal anddegrades range/sensitivity. In this four-detector system, the fourdetected RF output signals add as correlated voltages while the fournoise outputs add as uncorrelated power, improving the signal to noiseratio and providing low-cost detection systems capable of detecting lowor very low RF signals, whether reflected or remotely generated.

FIG. 5A is a circuit diagram of a single-diode detector 500 for use in adownconverting system according to an embodiment (see, e.g., FIG. 1,ref. num. 102). Other detector configurations are alternatively used. Adiode 502 is used as a peak detector to develop an output voltageV_(OUT) across a resistive-capacitive (“RC”) network 504 according to ahigh-frequency input voltage V_(IN) from a voltage source 506, andgenerates a detected output voltage approximately equal to the positivepeak input voltage minus the diode forward voltage drop of the detectordiode. The input voltage is, for example, the LO+RF voltage (see FIG. 1,ref. num. 110) developed by a local oscillator and an antenna, asdescribed below in reference to FIGS. 6A and 6B. In a particularembodiment, the value of the resistor 508 is selected to maximizedetected output voltage without adding excessive resistor noise and thecapacitor 510 is selected to provide low pass filtering and energystorage. The detector output connects to the high impedance input of anIF amplifier 510 according to an embodiment. This provides light loadingof the detector diode 502, which conducts over a narrow conduction angleof the high-frequency signal, compared to a conventional switchingmixer-based system where a mixer diode conducts over a relatively longportion of the high-frequency drive signal.

Switching mixer-based systems require relatively high LO power to drivethe mixer diodes. Using a diode as a detector, rather than as a mixer,allows operating the system with lower LO power, which results in lowertotal system power consumption and wider design choice in LO design. Alow LO power requirement at the detector diode also allows splitting theLO power to drive multiple detectors for LO AM noise canceling.Noise-canceling techniques according to embodiments providedownconverting systems with improved signal-to-noise performance overswitching mixers or single diode detecting mixers. Using multiple RFdetectors can further improve signal-to-noise performance.

FIG. 5B is a circuit diagram of a dual-diode detector 520 used in anembodiment. A voltage source V_(IN) 522 drives a first diode 524 toproduce a first output V_(OUT1) across a first RC network 526 and drivesa second diode 528 to produce a second output V_(OUT2) across a secondRC network 530. V_(OUT1) is essentially the positive peak voltage ofV_(IN) less the diode forward voltage drop, and V_(OUT2) is essentiallythe negative peak voltage of V_(IN) less the diode forward voltage drop.The detector outputs V_(OUT1), V_(OUT2) are provided to an algebraiccombining network 532 according to an embodiment.

FIG. 5C is a diagram of a diode multiplier circuit 540 used in anembodiment. The circuit 540 has four diodes 544, 546, 548, 560 connectedso as to provide voltage gain of approximately four over a single diodedetector. Each diode is driven by the AC input voltage and will chargeup the associated capacitor to which it connects as a peak detector.This detector (voltage multiplier) depends upon a low impedance driveand a high impedance load to deliver voltage gain. The diodes alternatein conduction such that the first and third diode conduct on thenegative half cycle and the second and fourth diodes conduct on thepositive half cycle of the input signal. The detected output is providedto a high-impedance circuit (e.g., an IF amplifier) 542 according to anembodiment.

FIG. 5D is a circuit diagram of a single-diode detector 560 in a shuntconfiguration used in an embodiment. The diode 562 works in cooperationwith a series capacitor 567 and a shunt resistor 564 to provide adetected voltage V_(DET) to the IF amplifier 566 according to anembodiment.

FIG. 6A is a plan view of a noise-canceling down-converting detector RFcoupler 600 for use in a field disturbance sensing system according toan embodiment. The RF coupler is fabricated as a transmission linehaving a selected characteristic impedance (e.g., 50 Ohms, 75 Ohms, or300 Ohms). The widths of the conductive traces are selected inaccordance with the thickness of the substrate (typically to a groundplane when a micro-strip transmission line is utilized) and with thedielectric constant of the substrate material and other characteristicsto obtain the desired characteristic impedance, as is well known in theart of RF circuits. For purposes of convenient discussion, the “length”of an RF structure, such as a segment of the ring coupler 602, will bereferred to in terms of the wavelength in which the system operates. Ina particular embodiment, micro-strip transmission lines are fabricatedon a dielectric substrate having relatively low dielectric loss(generally less than about 0.003 loss tangent at 10 GHz) and highdielectric constant (generally greater than about 2 at 10 GHz) such as aDUROID™ substrate, RODGERS RT™ 4350 or 4003 substrate available fromROGERS CORPORATION of Rogers, Conn., or ARLON-MED™ 25N, 25FR or AD350Asubstrate, available from ARLON-MED of Rancho Cucamonga, Calif., orTACONIC TLX™ or RF-35A™ substrate available from TACONIC of Petersburgh,N.Y., or ISOLA IS640™ available from ISOLA GROUP S.A.R.L. of Chandler,Ariz., which are poly(tetrafluoroethylene) (“TEFLON”)-based circuitboards with metal-foil traces. Micro-strip high-frequency transmissionstructures generally have a trace of a selected width separated from aconductive ground plane (typically, but not always, on the opposite sideof the substrate) by a known distance. Co-planar waveguide, stripline,single-sided stripline or co-axial transmission lines are used inalternative embodiments, or high-frequency transmission line types aremixed, such as using a micro-strip structure for one part of the systemand co-planar wave guide for another. Alternative embodiments use anepoxy-fiberglass substrate, such as an FR-4 or G-10 substrate, otherpolymer-fiber substrate, a ceramic (e.g., alumina or polysilicon)substrate or single-crystal (e.g., sapphire or silicon) substrate.

The noise-canceling down-converting detector RF coupler 600 uses a ringhybrid coupler 602 and two diode detectors 604, 606. The diode detectors604, 606 may be single or multiple diode detectors. The noise-cancelingdown-converting detector 600 is particularly desirable for coherentself-demodulated radar where the LO signal and the transmitted radarsignal 610 are at the same frequency with a constant phase difference.The RF signal 612 is the portion of the transmitted (LO) signal 610 thatis reflected by the target 614 with a frequency shift (i.e., Dopplershift) due to the target's velocity towards or away from the transmittedsignal source (transmitted signal 610). Thus the RF signal 612 isshifted by only a small amount from the LO frequency (compared with thecoupler bandwidth) and will have nearly the same wavelength as thattransmitted signal and thus similar phase shift, and the system can bedesigned for a known frequency, which is generally the transmittedfrequency, which in a particular embodiment is in one or more of the L-,S-, C-, X-, K-, Ku-, Ka-Band or other frequency. The details of theradar transmitting and receiving antennas are not shown in FIG. 6A, butthey may be implemented either with a common antenna and an LO and RFcombining network or be implemented with separate antennas for the LOand RF signals. The transmitted 610 and reflected 612 are routed throughan antenna coupler (see, e.g., FIG. 7A, ref. num. 708, FIG. 7B, ref.num. 757), which routes the reflected (RF) signal to the noise-cancelingdown-converting detector RF coupler 600.

The ring hybrid coupler 602 has four ports 616, 618, 620, 622. The ringhybrid coupler 602 splits the LO signal 615 arriving at the LO port 616into two equal signals 624, 626 and sends those LO signals to twodetector ports 618, 620 that are essentially identical. The LO signalstravel one quarter or an odd multiple of one quarter wavelengths of theLO signal from the LO port to either diode port (clockwise to diode port620 and counterclockwise to diode port 618). The RF port 622 is locatedon the ring hybrid coupler 602 a distance of an even multiple of onehalf wavelength clockwise from the LO port 616 and an odd multiple ofone half wavelength of the RF signal in the counter-clockwise directionaround the ring hybrid coupler 602.

The LO signals 624, 626 cancel at the RF port 622, as the minimum signalpath for 624 is ½λ (180 degrees), while the minimum signal path for 626is λ (360 degrees), thus the LO signals 624 and 626 arrive 180 degreesout of phase. Alternative embodiments use other multiples ofwavelengths. In an embodiment, the RF port 622 sits on the ring hybridcoupler 602 separated from one detector 604 by one quarter wavelength inthe counter-clockwise direction and 5/4λ in the clockwise direction, andseparated from the second detector 606 by three quarters of a wavelengthin either direction. Thus the RF signal arrives at each detector inphase from either direction around the ring. The closest distancebetween the two detector ports, 618,620 is ½λ. Thus, the signal from theRF port 622 splits and arrives at the two detectors 604, 606 with aone-hundred and eighty degree phase difference. The same phaserelationship can also be accomplished by scaling the ring by oddmultiples of ¼λ.

The RF signal 612 and LO signal 615 are not at exactly the samefrequency, but are separated by a very low frequency due to the Dopplershift of the moving target of the radar. This can be visualized as alarge LO signal adding with a smaller RF signal which appears nearlyidentical in frequency to the LO signal, but that moves slowly in phaseover time. For some cycles of the waveform, the RF and LO signals asseen by either of the detectors add in phase and increase the totalamplitude of the waveform. One half-cycle of the difference frequencylater, the RF and LO will add out of phase and decrease the totalamplitude of the waveform seen by that detector. This results in a lowfrequency output (i.e., baseband, or “audio”) from the diode detector atthe difference frequency between RF and LO (either LO−RF or RF−LO),which is the frequency that results from the change of the phase in thepath from the transmitted signal to the moving target and back. Thisconstant change in phase (for a constant relative velocity of thetarget) is indistinguishable from a change in frequency and isunderstood as the Doppler Effect.

The two detectors 604, 606 see the same phase of the signal from the LOport, but the small signal from the RF port adds to the amplitude of theLO signal at the first detector while it subtracts from the amplitude ofthe LO signal at the second detector. One half cycle of the differencefrequency later, the phase of the RF signal has changed by 180 degreesversus the LO signal, and the first detector which saw an addition ofthe RF magnitude and the LO magnitude will now see a subtraction of theRF magnitude from the LO magnitude. Thus if one detector sees a higheroutput, the other detector will see a lower output due to the same RFsignal. Coherent AM or AM noise of the LO signal will also show up as anaddition or subtraction of the signals at each detector which will addor subtract at each detector identically (i.e., in phase) such that thealgebraic combination of both detectors will see an increase in detectedRF output level, while detected LO AM noise at that combined output iscanceled.

An algebraic combining network (see FIG. 2A, ref. num. 218) takes thedifference between the two detector outputs and provides an IF signal.Thus, the AM noise of the LO signal is canceled, while the detectedamplitude of the RF signal shows up as the combination (sum) of the twodetected RF signals. The differential IF amplifier takes the differencebetween the detected RF amplitudes, which are nominally 180 degrees outof phase. This corresponds to adding another 180 degrees of phase to oneof the detected outputs and summing them, and is equivalent to addingthe magnitudes of the two detected RF signals, since a subtraction of anegative value is equivalent to addition of the magnitude of that value.

The differential detector and differential IF amplifier work together todeliver high sensitivity to the received RF signal while canceling theAM noise on the LO signal, which would otherwise limit the sensitivityof the downconverter. In a further embodiment, the differential IFamplifier provides adjustment of the amplitude of one IF signal comparedto the other IF signal (see, e.g., FIG. 3B, ref. num. 336) to furtherimprove noise-canceling performance of the system, particularly whenused in conjunction with a noise calibration technique in accordancewith an embodiment.

Some embodiments include noise calibration to correct for variations inthe detector gain and differences in signal path loss. Correct design ofthe RF transmission circuit and appropriate process control(repeatability) insures that the phase between LO signal portions andthe RF signal portions remain balanced. Even if the RF path has someimbalance, sufficient LO AM noise is canceled to result in only a slightmodification in the gain of the RF signals, while providing a largeincrease in the received signal to noise ratio.

The detectors 604, 606 do not operate like the diode switches used intypical mixers. The detectors 604, 606 can detect the input signalvoltage with higher gain (voltage output for voltage input) if they arenarrow bandwidth diode detectors, compared to the much wider bandwidthdiode switches used in switching mixer circuits. Narrow-bandwidth diodedetectors are easier to match than wider bandwidth diode switches,providing improved detection efficiencies and improved signal-to-noiseratio.

In an alternative embodiment, the RF port is connected to the couplersuch that the two detectors see the RF signal portions arrive in phasewith each other, but see the LO signal portions arrive 180 degrees outof phase with each other. The ring coupler geometry creates adifferential structure that cancels the AM noise on the LO signal. Thealgebraic combining network still subtracts the two detector outputs inorder to cancel the two detected LO AM noise portions, since the lowfrequency variation in AM noise will show up in phase at the two diodeoutputs regardless of the relative phase of the LO signals. Since the RFsignals add to both detectors, but the LO signals on those detectors are180 degrees out of phase with each other, the combination of LO and RFsignals create a low frequency variation on the output of the detectorsthat is 180 degrees out of phase and at the frequency difference betweenthe LO and RF signals. The algebraic combining network effectively addsthe amplitudes of the two detected RF signal portions. Thisimplementation may still require amplitude adjustment of one of thedetector outputs to obtain the maximum cancelation of the LO AM signalsor noise. Embodiments can incorporate single-diode or multiple-diodedetectors or use alternative detector methods.

The detector outputs are coupled to a summing network (not shown, seeFIG. 7B, ref. num. 768). The first stage of the summing network is an IFamplifier that receives the detector outputs from the detectors 604, 606of FIG. 6A. This amplifier is a differential amplifier formed from anemitter coupled transistor pair. The gain may be changed by changing thebias current of the two transistors forming the differential amplifier.The outputs of the first stage of the IF amplifier for one detector areconnected to a summing amplifier (e.g., an op amp), the positive outputcoupled to the input of an amplifier generating non-inverting gain andthe negative output coupled to the input of an amplifier generatinginverting gain. Because this network takes the difference between thetwo detectors, the outputs of the first stage of the second detector arecoupled to the opposite polarity of the amplifier inputs, i.e., thepositive output of the second detector runs to a port generatinginverting gain, and the negative output of the second detector runs to aport generating non-inverting gain. This configuration generates adifference between the two detector outputs and cancels common modenoise variations in the supply or gain control voltages for the firststages of the IF amplifier. The differential character of thedifferential noise canceling detector preserves noise cancelation at allelements of the receiver chain up to the amplifier output. In a furtherembodiment, the amplifier is replaced by a differential amplifier withdifferential outputs, which further reduces sensitivity to common modeinterference.

FIG. 6B is a plan view of a single-diode detector circuit 630 accordingto an embodiment. The detector circuit is used as the first or seconddetector 604, 606 of FIG. 6A, for example. A diode 632 is matched to thetransmission line impedance of the ring coupler (see, e.g., FIG. 6A,ref. num. 602) with a matching structure 634. The diode 632 is a diodechip that is that is mounted on a pad 636 of conductive foil defined onthe surface of the circuit substrate, and is connected to the matchingstructure 634 with a bond wire or other suitable connector 638, which isrepresented as an inductor. Packaged diodes are used in alternativeembodiments.

The diode uses a DC path to develop a current and provide a detectedoutput (V_(OUT)) from the incident high-frequency power (i.e., LO and RFsignals). A network with fan lines 642, 646 on opposite ends of ahigh-impedance transmission line 644 provides both a high impedance atthe LO frequency and a DC path for diode current, as is well known inthe art of high-frequency hybrid microcircuit design. Any of severalmatching and bias structures are suitable in various embodiments, andsome embodiments may utilize different matching and biasing techniqueson different detector circuits. Accordingly, the detector circuit 630 ismerely exemplary. Alternative embodiments use detector circuits with oneor more diodes in a shunt configuration. It is generally desirable toprovide a DC connection to the output of the diode without loading theresonance of the diode inductance and fan line 642 capacitance. Manymixer circuits operate in a relatively low characteristic impedancesystem, such as a 50-ohm system. Detectors in some embodiments operatein circuits with much higher impedance, which avoids loading theresonance and delivers higher voltage gain from the detector.

A second fan line 646 connects to a network 650 that has a seriesresistor 652, shunt capacitor 654 and a shunt resistor 656. The seriesresistor 652 should have a resistance large enough to provide isolationbetween the fan 646 and the following shunt capacitor 654, yet have aresistance sufficiently lower than the shunt resistor 656 to ground, soas not to unduly attenuate the output voltage _(VOUT). For example, witha shunt resistor 656 in the range of 1 to 5 K ohms, the series resistor652 is between about 20 ohms and about 200 ohms.

The shut capacitor 654 is chosen to have a self-resonant frequency abovethe highest expected IF frequency, and below the LO frequency. In aparticular embodiment with an LO frequency of about 24 GHz, the shuntcapacitor is chosen to have a self-resonant frequency not greater thanabout two GHz. This provides some immunity from adjacent signal sources,such as other radar units or communications devices, from affecting theoutput voltage V_(OUT). The shunt capacitor 654 and shunt resistor 656in conjunction with the resistance and capacitance presented by thefollowing amplifier connected to Vout will determine the IF bandwidth,which must be set large enough to enable reception of the highestfrequency IF signal expected to be received. The shortened fan lines642, 646 and series inductances of the diode and package, line 644, andother leads, provide attenuation at LO and RF frequencies. Several othernetworks are alternatively used, as would be appreciated by one of skillin the art.

Transmission lines 658, 660, 662 are used in the matching structure 634to match the impedance of the diode/fan line resonance to the systemimpedance of the ring coupler. Other transmission line matchingstructures are alternatively used. For example, an alternative designuses a single one quarter wavelength long transmission line withimpedance equal to the geometric mean of the source and load impedances.In an alternative embodiment, discrete components are used in a matchingcircuit.

FIG. 6C is a plan view of a dual-diode detector circuit 660 according toan embodiment. Two diodes 662, 664 are connected in series. In aparticular embodiment, the diodes are made in a single package 666 toreduce stray capacitance and inductance, and the package of two diodesis referred to as a dual diode. Some embodiments use stacked diodes. Thecommon junction 668 of the two diodes connects through a wide/narrowmatching structure (see, e.g., FIG. 6B, ref. num. 634) to the detectorinput port (e.g., FIG. 6B, ref. num. 618 or 620), as discussed above inreference to FIG. 6B. The other terminals of the dual diodes are eachconnect to a shortened fan line 670, 672 emulating a capacitance to tuneout the diode and package lead inductances and resonate at the LOfrequency. Each diode and shortened fan line junction connects to acircuit presenting a high impedance load at the LO frequency, butconnects to the diode at DC to provide a detector output and a DCcurrent path. This circuit may take the form of the narrow line and fanline plus series resistor and shunt resistor and capacitor as describedabove in reference to FIG. 6B. An alternative embodiment takes the formof a high impedance network at the LO frequency such as an RF choke,which provides a high impedance. One diode of the dual diode packagedetects the positive peaks of the detector input signal and the otherdiode of the dual diode package detects the negative peaks of thedetector input signal (see, e.g., FIG. 5B and associated description).

The outputs V_(OUT1), V_(OUT2) of the two diodes are connected to theinputs of a differential amplifier 676. The output 678 of thedifferential amplifier 676 is connected to the input of a differentialIF amplifier 682. The dual diode detector circuit 660 operates similarlyto the single diode detector circuit of FIG. 6B, but typically deliverssignificantly higher output voltage than a single diode detector,depending upon the losses in the matching networks and straycapacitances and inductances.

FIG. 7A is a diagram of a portion of a CW radar system 700 according toan embodiment. A first ring coupler 708 routes the LO signal from alocal oscillator 720 to an antenna 722 and routes the RF signal from anantenna 722, which in a particular embodiment is a patch antennafabricated on the same substrate 723 as the ring couplers 702, 708. Thesubstrate 723 is commonly known as a “printed circuit board”. A singleantenna 722 is used in the CW radar system 700 to both transmit the LOsignal (see, FIG. 6A, ref. num. 610) and to receive the reflected RFsignal. Alternative embodiments use two antennas, one for transmittingand one for receiving allowing for additional receiver gain for bettersensitivity and range. The gain increases because the received signal atthe antenna may connect directly to the NCDD input as opposed to anantenna coupler, which typically introduces about 3 dB of loss. Theantenna is designed to operate at the LO frequency, which in aparticular embodiment is about 24 GHz. The RF and LO are essentially thesame frequency in a CW Doppler radar system. A second ring coupler 702is configured substantially as described above in reference to FIG. 6A,and includes two detectors 704, 706 substantially as described above inreference to FIG. 6B. Alternatively, multiple-diode detectors are usedfor one or both of the detectors 704, 706.

Each ring is optimized to maintain to best performance of the overallsystem. In a particular embodiment, the local oscillator 720 is adielectric resonator oscillator (“DRO”, also known as dielectricresonator stabilized oscillators (“DSOs”)). DROs are low-cost, compact,and consume relatively little power consumption; however, DROs oftenhave too much AM noise to be used in conventional diode detector CWradar systems without noise cancelling. Embodiments of the inventioncanceling LO AM noise allow DROs to be used in many different radarapplications, including radar applications requiring low powerconsumption, long range, or measurement accuracy. Alternativeembodiments use a transmission line resonator oscillator or otheroscillator.

The first ring coupler 708 receives the LO signal at a first port (“LOinput port”) 724 and distributes the LO signal to an antenna port 726for transmission to the target (not shown) and an LO port 728. Theportion of the LO signal coupled to the LO port 728 is transmittedthrough a transmission line 730 to deliver the LO signal to the LO port616 (“of the second ring coupler 702. The portion of the LO/RF signalcoupled to the RF port 732, which is directly across from the LO port728 and isolated from LO signal is coupled to the RF port 622 of thefirst ring coupler 702. Other details of the radar system 700 are wellknown in the art of RF circuit design and are omitted for clarity ofillustration.

The LO input port 724 is separated from the LO output port 728 by ¼λ,from the antenna port 726 by ¼λ and from the RF output port 732 by ½λ CWand 1λ CCW for a difference of 180 degrees. This allows the LO signal tosplit into 2 paths, to the antenna and to the differential detector. Thesecond ring hybrid coupler sends the LO signal to the two detectordiodes nominally in phase and the RF input signal from the antennanominally 180 degrees out of phase at the two diode detectors. In aparticular embodiment, the LO 720 is fabricated on the opposite side(“second side”) of the substrate 723 from the side (“first side”) of thesubstrate that the ring couplers 702, 708 and antenna 722 are fabricatedon. A ground plane 734 on the first side overlies the LO circuit on thesecond side (not shown), and the LO signal is brought from the secondside, where it is generated, to the first side through a plated via 736.A ground plane on the second side (not shown) underlies the antenna 722and generally the RF circuitry on the first side, as is known in the artof RF microstrip design.

FIG. 7B is a diagram of a field disturbance measuring system 750according to an embodiment. In a particular embodiment, the system 750is an integrated CW radar system. The system 750 includes an antenna 752that transmits a signal (LO signal) 754 generated by an oscillatorcircuit 756 at a target 755. The target 755 is shown in dashed linesbecause the target is not part of the system. The system measures thespeed at which the target is moving relative to the system, if any. Thetarget may be stationary while the system is moving, the target may bemoving while the system is stationary, or both target and system may besimultaneously moving relative to the general landscape or otherreference frame.

The antenna 752 and receives a signal (RF signal) 758 reflected off thetarget 755. The reflected signal is combined with the LO signal in acombining network 757 (see, e.g., FIGS. 1-2B and associated WrittenDescription) and provided to a first detector 760 and to a seconddetector 762 that operate as differential detectors to cancel AM noise(see, e.g., FIGS. 3A-4B and associated Written Description). In analternative embodiment, a first antenna is used to transmit the LOsignal and a second antenna is used to receive the reflected signal. Anamplifier (preamplifier) is optionally placed after the antenna in thereflected signal path.

The outputs of the first and second detectors 760, 762 are provided toan algebraic summing network 768. Differential outputs of the summingnetwork 768 are coupled to an amplifier and filter 770, which convertthe differential inputs to a single-ended signal 771. Ananalog-to-digital converter 772 converts the signal, representing thedetected voltage, into a digital value 773 that is processed by acontroller 774, which renders the measured speed (between the target andsystem) to an electronic display screen 776, such as a liquid crystaldisplay screen. The system 750 optionally includes a user interface 778for communicating information such as the measured speed to an externaldevice, accepting a trigger to start the radar speed measurementprocess, modifying the function of the radar to report the speed inalternative units or optimizing the radar speed measurement process toreport the speed of a particular type of object (i.e. optimized formeasuring a vehicle or a baseball) or report the measured speed within aparticular limit of high and low speeds. In an alternative embodiment, asystem does not include an integrated display, and the controllerprovides relative speed data to a device (not shown) outside of thesystem.

In a particular embodiment, the system 750 is a portable system poweredby a battery(s) 784. In a further embodiment, the portable system isintended to be operated as a hand-held system. In an alternativeembodiment, the battery is not included in the system, and power issupplied from an external source, such as an external battery (e.g., avehicle battery) or mains power (e.g. from a transformer connected tothe mains power). A power supply circuit 786 provides voltage regulationand similar functions to generate the appropriate voltages and supplycurrents to power the system components. Individual power lines to eachpowered component are omitted for simplicity and clarity ofillustration. In a particular embodiment, operation of the powersupplies are monitored by the controller 774 and A/D converter 772, andadjusted or turned off and on, if necessary, through a power supplycontrol line or lines 787.

The controller 774 provides an amplitude modulation control signal 788to the oscillator circuit 756, which creates amplitude modulation of theoscillator 756 to enable calibration of the NCDD to minimize thedetected level of AM noise of the oscillator 756 by the NCDD. In aparticular embodiment, the controller 774 includes memory, such ascalibration tables, or the system 750 includes separate memory (notshown) that cooperates with the controller. In a particular embodiment,the controller 774 includes a signal processing block and an optionalautomatic calibration block that works in cooperation with an AM signalgenerator (i.e., a signal on the AM control line 788 that modulates theoscillator 756 in a known fashion). In a particular embodiment, thesystem is integrated in a housing 790 containing the other systemelements for hand-held application. In a particular embodiment, thehousing 790 is a rectangular plastic housing having approximatedimensions of 2.25 inches by 4.5 inches by one inch. Alternative systemsare provided as original-equipment manufacturer (“OEM”) systems, andincorporated into other products wherein the housing may be omitted.

FIG. 8A is flow chart of a method of down-converting 800 according to anembodiment. A high-frequency signal (e.g., the LO signal) is generated(step 802). The high-frequency signal is provided to an antenna, a firstdetector and a second detector (step 804). The antenna receives areflected signal (e.g., the RF signal) from a target (step 806), and thereflected signal is provided to at least the first detector (step 808).The first detector converts the reflected signal and high-frequencysignal to a first detected output (step 810), and the second detectorconcurrently converts at least the high-frequency signal to a seconddetected output (step 812). The first and second detected outputs arealgebraically combined (e.g., subtracted from each other) so as tocancel the detected high-frequency signals, including AM noise on thedetected high-frequency signals (step 814). The LO AM noise iscorrelated between the detectors and their outputs will increase forhigher LO power or decrease for lower LO power on both detectors,regardless of the phasing of the LO signals sent to the detectors, thusdetector output algebraic combiners will subtract one detector outputfrom the other output. Embodiments of the method of FIG. 8A are used inmotion sensing systems, such as door openers, distance ranging systems,automotive speed or range sensors or low IF receivers. In a particularembodiment, the combined detected reflected signal is processed toderive a relative speed between the radar system and the moving target(step 816).

In a particular embodiment, the second detector detects the same phaseof the RF signal as the first detector, and the opposite phase (i.e.,180 degrees out of phase) of the LO signal. One of the detected RFsignals is inverted at the detector output and subtracted from the otherdetected RF signal, while the detected LO AM signals cancel due to thesame subtraction. In an alternative embodiment, the second detectordetects the opposite phase of the RF signal as the first detector, andthe same phase of the LO AM signal. The output of one of the detectoroutputs is subtracted from the other to add the detected (downconverted)out of phase RF signals and cancel the detected in phase (demodulated)LO AM signals. In both cases the demodulated amplitude modulation of theLO signals comes out of the detectors in phase at the detector outputsand the downconverted RF signal comes out of the detectors out of phaseat the detector outputs.

FIG. 8B is a flow chart of a method of sensing an electro-magnetic fielddisturbance 820 according to an embodiment. A high-frequency signal(e.g., the LO signal) is generated (step 802). The high-frequency signalis provided to an antenna, a first detector and a second detector (step804). The antenna receives a reflected signal (e.g., the RF signal) froma target (step 806). A reflected signal is provided to the firstdetector and an inverse reflected signal is provided to the seconddetector (step 822). The first detector converts the reflected signaland high-frequency signal to a first detected output (step 810), and thesecond detector concurrently converts the inverse reflected signal andhigh-frequency signal to a second detected output (step 824). In aparticular embodiment, the first detected output is a detected LO signaland a detected RF signal, and the second detected output is essentiallythe first detected LO signal and a negative detected RF signal.

The first detected output is provided to a first algebraic combiningnetwork and the second detected output is concurrently provided to asecond algebraic combining network (step 826). The first algebraiccombining network produces a first differential signal and an inversefirst differential signal and the second algebraic combining networkproduces a second differential signal and an inverse second differentialsignal (step 832). The first differential signal is provided to apositive input (i.e., non-inverting input) of a third algebraiccombining network, the inverse first differential signal is provided toa negative input (i.e., inverting input) of the third algebraiccombining network, the second differential signal is provided to asecond negative input of the third algebraic combining network and theinverse second differential signal is provided to a second positiveinput of the third algebraic combining network. The third algebraiccombining network produces an output (e.g., the IF output) (step 830).In a further embodiment, the combined detected reflected signal isprocessed to derive a relative speed between the radar system and thetarget (step 832).

In a further embodiment, gain (including negative gain, which is alsoknown as attenuation) is applied to the output of the second algebraiccombining network (step 834) to match the detected LO signal from thefirst detector to the detected LO signal from the second detector so asto cancel AM noise detected on the LO signals.

FIG. 9A is a flow chart of a method of calibrating noise 900 in a fielddisturbance sensing system according to an embodiment. A high-frequencysignal (e.g., the LO signal) is applied to an antenna, a first detectorand a second detector of the field disturbance sensing system (step902). The first and second detectors are arranged as noise cancelingdetectors (see, e.g, FIGS. 1-3B). The antenna is shielded so as to notreceive reflections from moving objects or other external radiation(step 904). Shielding can be performed before or after thehigh-frequency signal is applied. Shielding can be performed in avariety of ways, such as pointing the antenna into an open box linedwith radio frequency absorbing material or pointing the antenna towardsan area with no target to create a reflection.

Gain adjustments are stepped through a selected range of settings (see,e.g., FIG. 3B, ref. no. 336) and the output level (e.g., IF level 338 orother suitable signal level), which indicates the differential noise, isrecorded (step 906). The gain setting associated with the lowestdetected AM noise level is identified (step 908), and then saved (step910). During operation, the gain setting associated with the lowestdetected AM noise level is applied, and a noise canceling fielddisturbance measurement is made (step 912).

Gain is adjusted in various ways in alternative embodiments. Forexample, an adjustable gain element is included in one of the detectedsignal paths to increase or decrease the gain of that detected signalrelative to another detected signal. Alternatively, the bias ofelements, such as the bias level to one or more transistors, is changed.As the gain in one of the detector paths changes, the level of noisewill fall to a minimum value. By observing the noise level for each biascontrol step, one can identify the gain setting where the noise beginsto increase as the bias control leaves that optimum area (e.g.,increasing or decreasing bias control voltage above or below the biascontrol voltage at the minimum noise condition). In a particularembodiment, a threshold is defined at a selected excursion from theminimum point. The gain settings at which the noise equals the thresholdis determined (i.e., the minimum noise will be between the two thresholdsettings, but may be relatively “flat”, making a direct measurement ofthe minimum noise point less precise). The optimum point for best LOnoise rejection is set by choosing the gain setting between the gainsettings for the two thresholds identified above. In embodiments thatuse more than two detectors (see, e.g., FIG. 4), the gain of a combineddetector signal (e.g., the output of network 402 or network 404) may beadjusted so as to achieve minimum AM noise.

FIG. 9B is a flow chart of a method of calibrating noise 920 in a fielddisturbance sensing system with an AM generator according to anembodiment. A high-frequency signal (e.g., the LO signal) is applied toan antenna, a first detector and a second detector of the fielddisturbance sensing system (step 922). The first and second detectorsare arranged as noise canceling detectors (see, e.g, FIGS. 1-3B). Theantenna is shielded so as to not receive reflections from moving objectsor other external radiation (step 924). Shielding can be performedbefore or after the high-frequency signal is applied, and before orafter the calibration signal is applied, as long as the shielding is inplace before the measurement sequence begins.

A calibration signal that mimics AM noise of the LO, or a signal at anoperating frequency (e.g., the LO frequency plus sidebands) is appliedto a first detector and to a second detector of a noise-cancelingdown-converting detector system (see, e.g., FIGS. 1, 2A, 3B). In aparticular embodiment, the LO is modulated with an AM signal generatedby the system (see, e.g., FIG. 7B, ref. nums. 756, 788, 744) (step 926).Gain adjustments are stepped through a selected range of settings andthe output level (e.g., IF level 338 or other suitable signal level),which indicates the differential noise, is recorded (step 928). The gainsetting associated with the lowest detected AM signal (AM noise) levelis identified (step 930), and the saved (step 932). During operation,the gain setting associated with the lowest detected AM signal outputlevel is applied (step 934), and a noise canceling field disturbancemeasurement is made (step 936).

While the invention has been described with reference to a preferredembodiment or embodiments, it will be understood by those skilled in theart that various changes may be made and equivalents may be substitutedfor elements thereof without departing from the scope of the invention.In addition, many modifications may be made to adapt a particularsituation or material to the teachings of the invention withoutdeparting from the essential scope thereof. Therefore, it is intendedthat the invention not be limited to the particular embodiment disclosedas the best mode contemplated for carrying out this invention, but thatthe invention will include all embodiments falling within the scope ofthe appended claims.

1. A detector system comprising: a local oscillator port; a radiofrequency port; an oscillator providing a local oscillator signal havingamplitude-modulated (“AM”) noise coupled to the local oscillator port; afirst detector configured to detect at least the local oscillator signalto produce a first detected signal having at least a first detected AMnoise signal component and a demodulated signal component at a firstphase; a second detector configured to detect a second high-frequencysignal having the AM noise to produce a second detected signal having atleast a second detected AM noise signal component at a second phase; analgebraic combining network combining the first detected signal and thesecond detected signal to produce an output signal including thedemodulated signal component at an output; and a phase delay disposedbetween one of the local oscillator port and the radio frequency portand the output of the algebraic combining network calibrated to offsetthe first phase from the second phase to improve signal-to-noise ratioat the output.
 2. The detector system of claim 1 wherein the localoscillator signal is at a selected local oscillator signal level and thephase delay is calibrated at the selected local oscillator signal level.3. The detector system of claim 1 further comprising a high-frequencycombining network, wherein the phase delay is disposed in a radiofrequency path of the high-frequency combining network.
 4. The detectorsystem of claim 1 further comprising a high-frequency combining network,wherein the phase delay is disposed in a local oscillator path of ahigh-frequency combining network.
 5. The detector system of claim 1wherein the phase delay is disposed between the first detector and theoutput of the algebraic combining network.
 6. The detector system ofclaim 1 wherein the phase delay is disposed between the second detectorand the output of the algebraic combining network.
 7. The detectorsystem of claim 1 further comprising a second phase delay calibrated toimprove signal-to-noise ratio at the output